Direct-conversion transmitting circuit and integrated transmitting/receiving circuit

ABSTRACT

A transmitter is provided which includes a transmitting circuit that does not require a high-performance low noise VCO restricting cost reduction thereof and that can reduce the number of parts without requiring an RF filter. A direct conversion that does not require a transmission VCO is applied to the transmitting circuit. In order to achieve noise reduction in a receiving band, low-pass filters are provided at IQ input sections of a modulator that converts IQ signals into RF signals. In comparison with a conventional transmitter using offset PLL, an external VCO required in addition to an RF integrated circuit, a power amplifier, and a front end circuit is reduced. Even in current transistor performance, by using a filter having rapid waveform characteristics such as a SAW more inexpensive than the VCO, or the like, it is possible to provide a GSM/GSM 1800/GSM1900 triple band transmitter.

This is a continuation application of U.S. application Ser. No.10/073,029, filed Feb. 12, 2002, the entire disclosure of which ishereby incorporated by reference.

BACKGROUND OF THE INVENTION

The present invention relates to mobile communication equipment, andparticularly to a direct-conversion transmitting circuit suitable forlarge scale integration and to an integrated transmitting/receivingcircuit using the same.

According to rapid spread of the mobile communication equipment,requests for miniaturization and lower cost thereof have increased.Because of this, it is expected to apply a voltage control typeoscillator (VCO), or an integrated circuit whose filter number isreduced and whose integration is enhanced. What is given as oneconventional example of transmitting equipment is “RF Circuits Techniqueof Dual-Band Transceiver IC for GSM and DCS1800 applications” publishedin pages 278 to 281 of manuscripts for IEEE 25th European Solid-StateCircuits Conference on 1999 by K. Takikawa et al.

As an important item on a transmitting circuit design, reduction ofnoise leakage into receiving frequency band has been given. FIG. 2 showsa relationship between transmission power defined by specifications forEuropean cellular phones (GSM) and noise generated in receiving band. Asindicated by an allowed output spur level 202 in a GSM receiving band,it is required that noise in receiving a band 204 at slightly 20 MHzdistant from an upper limit of a transmission band 203 is suppressed upto −79 dBm/100 kHz (−129 dBm/Hz) or less relative to a maximum outputpower of 33 dBm in a GSM output signal 201. That is, a differencebetween a transmission signal and a noise level is required to be −112dBc or more. If a band pass filter or the like is applied to an outputportion of a power amplifier, the above-mentioned specification can beachieved. However, decrease in efficiency thereof is generated due toinfluence on losses of the filter. Thus, an offset PLL is applied as aconstitution using no filter.

FIG. 18 is a circuit constitution diagram showing a transmitter to whichthe offset PLL is applied. The transmitter is composed of an IF signalgenerating section 1815 and a PLL section 1814. First, an operation ofthe IF section will be described here. I and Q signals each having aband of 200 kHz are input. These input signals are mixed in intermediatefrequency (IF) local signals 1812 and 1813 each having a phasedifference of 90° and mixers 1808 and 1809, respectively. In this case,the local signals 1812 and 1813 are obtained by phase-shifting an outputof the VCO 1806 by a 90° phase shifter 1807. By adding outputs of themixers 1808 and 1809, the input signals are converted into a GMSK(Gaussian Minimum Shift Keying) modulation signal of an IF frequency(270 MHz), respectively. The GMSK modulation signal is a modulationsignal adopted in the GSM system, and has signal information only in aphase, and is constant in amplitude. In order to provide a sufficientamplitude to a phase comparator 1802 located at a downstream side of thecircuit, the IF signal is amplified at an amplifier 1810. Afterharmonics generated by the mixers 1808 and 1809 and the amplifier 1810are removed by a low-pass filter 1811, the IF signal is input to thephase comparator 1802 of the a PLL section 1814.

The PLL section 1814 is characterized by including a mixer 1801, andconverts a frequency of an output signal of the VCO 1800 operated by anRF frequency, into an IF frequency f_(IF) (270 MHz), by means of a mixer1801, and outputs amounts of error between the IF signal and the outputsignal of the VCO 1800 by means of the phase comparator 1802. Afrequency of the output error signal is lowered up to a baseband signalband that is the same as the I and Q input signals. High frequency noiseof the error signal is suppressed by the low-pass filter 1803. A cutofffrequency of a PLL closed loop of a filter, the PLL closed loop which isdenoted by reference numeral 1816, is about 1.6 MHz in a signal band of200 kHz, and a noise of 20 MHz is greatly suppressed. Because of this,the noise generated in band which is a 20 MHz distant from an outputsignal of the VCO 1800 is greatly suppressed. Therefore, even if anoutput of the VCO 1800 is directly connected to a power amplifier PA, itis possible to suppress noises of receiving band up to −79 dBm/100 kHz(−129 dBm/Hz) or less without newly connecting a filter to an RF signal.

In a transmitter using the offset PLL, although a portion 1817 enclosedin solid line shown in FIG. 18 is integrated, the VCO 1800 is anexternal part because low noise characteristics are required. However,if the offset PLL is used, an external filter for high frequency is notrequired. Therefore, it is possible to be widely applied as atransmitter with high efficiency.

SUMMARY OF THE INVENTION

As described previously, a conventional example applying an offset PLLhas been used as a transmitter because requiring no external filter.However, in the transmitter applying the offset PLL, there has been thelimit of cost reduction because an external VCO with low noise isrequired.

An object of the present invention is to provide a direct-conversiontransmitting circuit that does not require a low noise VCO having highperformance and restricting the cost reduction in order to achievereduction of the number of parts thereof, and that does not require anexpensive and external high frequency filter such as a surface acousticwave (SAW) filter or the like.

An object of the present invention is also to provide atransmitter/receiver using a direct-conversion transmitting circuit.

In order to solve problems described above, a transmitting circuitaccording to the present invention has an element of a circuit that usesa direct-conversion requiring no transmission VCO, and that provides alow-pass filter with each of I and Q (hereinafter, referred to as IQ)input portions of a modulator for converting IQ signals to a RF signalin order to achieve noise reduction in receiving band. An integratedtransmitting/receiving circuit according to the present invention usesthis direct-conversion transmitting circuit in a transmitting circuitsection thereof. Concrete descriptions will be made in the followingembodiments.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a configurational view showing an embodiment that is adirect-conversion transmitting circuit according to the presentinvention.

FIG. 2 is a view showing a relationship between transmission powerdefined by a GSM specification and noises in receiving band.

FIG. 3 is a view showing a relationship between transmission powerdefined by a GSM1800 specification and noises in receiving band.

FIG. 4 is a view showing noise generation factors of a direct-conversiontransmitter.

FIG. 5 is a trial manufacture circuit diagram used for checking effectsof the present invention.

FIG. 6 is a characteristic-wiring diagram between an input signal leveland a noise level showing measurement results of the circuit shown inFIG. 5.

FIG. 7 is a characteristic-wiring diagram between a cutoff frequency anda phase precision of a filter showing the measurement result of thecircuit shown in FIG. 5.

FIG. 8 is a level diagram showing a transmitter of a first embodimentthat is the present invention used as an example of a GSM1800.

FIG. 9A is a view showing amplitude response of a first and secondfilters.

FIG. 9B is a view showing group delay response of a first and secondfilters.

FIG. 10 is a configurational view of a mixer circuit using a secondfilter showing a second embodiment of the present invention.

FIG. 11 is a view showing a third embodiment of the present invention,and is a circuit configurational view provided with a function ofcorrecting a DC offset of a mixer constituting an modulator.

FIG. 12 is a view showing a fourth embodiment of the present invention,and is a configurational view reducing the use number of AD convertersfor DC offset correction.

FIG. 13 is a view showing a fifth embodiment of the present invention,and is a circuit diagram of an important part of a configuration using asecond order filter as input of a mixer section which constitutes amodulation section.

FIG. 14 is a view showing a sixth embodiment of the present invention,and is a circuit diagram of an important part of a configuration using aMOSFET as a second order filter provided for an input of a mixer sectionwhich constitutes a modulation section.

FIG. 15 is a view showing a seventh embodiment of the present invention,and is a circuit diagram of an important part showing a configuration ofa transmitting/receiving integrally integrated circuit that correspondsto a GSM, GSM1800, and GSM 1900.

FIG. 16 is a view showing an eight embodiment of the present invention,and is a configurational view showing the case of having doubled anoperating frequency of an oscillator in the circuit of FIG. 15.

FIG. 17 is a view showing a ninth embodiment of the present invention,and is a circuit diagram of an important part showing a configuration ofa transmitting/receiving integrally integrated circuit that correspondsto a GSM850, GSM, GSM1800, and GSM1900.

FIG. 18 is a circuit diagram of an important part showing aconfiguration of a conventional transmitter which uses an offset PLLcircuit.

FIG. 19 is a level diagram of a transmitter according to a firstembodiment of the present invention that uses a GSM as an example.

FIG. 20 is views showing an input waveform, a two-frequency divisionwaveform and a four-frequency division waveform of two frequencydividers that are used for generating a local signal in the constitutionof FIG. 16

DESCRIPTION OF THE PREFERRED EMBODIMENTS

Preferred embodiments of a direct-conversion transmitting circuit and anintegrated transmitting/receiving circuit using the same according tothe present invention will be described in detail below, with referenceto the accompanying drawings.

First Embodiment

A first embodiment of the present invention will be described withreference to FIGS. 1 to 8 and FIG. 19. In this embodiment, similarly toa conventional example, European cellular phone GSM (900 MHz band) andGSM1800 (1800 MHz band) are targeted as applications. Although there hasalready been given in the description of the conventional technique, anoise level from a transmitter within receiving band that the GSMspecification should satisfy has been shown in FIG. 2. FIG. 3 showsconditions that the GSM1800 should satisfy. A transmission band (TXBand) 303 is within the range of 1710 to 1785 MHz, and a receiving band(RX Band) 304 is within a range of 1805 to 1880 MHz. Therefore,similarly to the case of the GSM, there is an interval of 20 MHz. It isrequired to suppress a noise of −71 dBm/100 kHz (−121 dBm/Hz) or less inorder to satisfy an allowed spur level 302 in receiving band relative toa maximum transmission power of 30 dBm in an output signal 301 of theGSM1800. Thus, a level difference between the maximum transmission power301 of the transmission signal and the noise in receiving band is −101dBc as shown in FIG. 3. As compared with the GSM, this specification isreduced by 11 dB.

Here, problems, which arise in a direct-conversion transmitting circuitand should be solved, will be clarified with reference to FIG. 4.

A direct-up-conversion transmitter has the same circuit constitution asthe conventional IF section 1815 shown in FIG. 18, but a modulationmethod for directly generating a RF modulation signal is appliedtherein. Relative to the direct conversion, a direct-up-conversion meansthe case of executing a frequency conversion that straight increases afrequency from a baseband to a transmission frequency band, and adirect-down-conversion means the case of executing a frequencyconversion that straight decreases a frequency from a receivingfrequency band to a baseband.

In FIG. 4, AD converters (ADC) 402 and 403 generating respective IQsignals are also shown in addition to a modulation 106 constituted bymixers 101 and 102. As shown in a relation between frequency “f” andoutput “p”, which is illustrated in a rectangle 410 and displaysfrequency characteristics of a signal/noise level of an output sectionin the AD converter 402, an output section of the AD converter 402includes a signal body 411, a turn noise 412, and a thermal noise 413which a circuit thereof has. An output section of the AD converter 403also has the same characteristics as that of the AD converter 402.

In order to suppress the turn noise, low-pass filters 404 and 405 areinstalled at each output section of the AD converters. An output of thelow-pass filter 404 is shown in a rectangle denoted by reference number414, and suppresses a signal equal to or more than a cutoff frequencyf_(CUTOFF) of the low-pass filter 404, and includes a signal 415 and anoise 416 which is equal to or less than the cutoff frequency. Thelow-pass filter 405 also has the same output as the low-pass filter 404.Output signals of the low-pass filters 404 and 405 are applied to an Iinput 108 and a Q input 109 of the transmitting circuit, respectively.An optimal output signal amplitude of the AD converter is about 2 Vpp innormal differential. On the other hand, an optimal input level of themixer depends on a circuit constitution and, for example, is 0.8 Vpp,and so is different from the output signal level of the AD converter.Besides, since optimal bias levels thereof are also different from eachother, attenuators 103 and 104 each including a shift function of a biaslevel are required.

Since generating noises, an output of the attenuator 103 includes asignal 419, noise 420 equal to or less than the cutout frequency of thelow-pass filter, and further a noise 421 generated by the attenuator asshown in a rectangle 418. The attenuator 104 also has the same output asthe attenuator 103.

Each noise of the attenuators 103 and 104 extends within a wide band. Inthe mixers 101 and 102, the IQ signals containing noises are convertedinto RF frequencies that each regard a carrier frequency “fc” as acenter. Local signals having the same frequency as the carrierfrequencies whose a phase difference is 90° and which have the samefrequencies are applied to two mixers 101 and 102. Each local signal isgenerated by a phase shifter 100 in accordance with an output of anoscillator 105. As typical phase shifters, there are two types of oneusing a CR filter and one using a frequency divider.. However, in thecase of one using the frequency divider, an oscillation frequency of theoscillator 105 is twice as high as the carrier frequency.

An output of the modulator 106 composed of the mixers 101 and 102becomes a signal modulated in both sides by regarding the carrierfrequency “fc” as a center, as indicated in rectangle 422. Thismodulation signal includes a modulator output signal 424, and a thermalnoise 425 caused by the AD converter for a modulator output, and a noise426 caused by the attenuator for a modulator output. In particular, thenoise 421 of the attenuator extends within a wide band, and themodulated noise also exists as the noise 426 caused by the attenuator ina wide band.

A modulation signal is further amplified by a driver amplifier 107, andis output via an output terminal 132. This signal includes a wide bandnoise therein. Thus, in order to reduce a noise within a receiving bandwhich is 20 MHz distant from a transmission band, a high frequency (RF)filter 430 which has a rapid waveform characteristic and which applies aSAW (surface acoustic wave) device, dielectric resonator, or the like isrequired.

In order to eliminate the high frequency filter, it is required toreduce a wide band noise each generated by the attenuators 103 and 104.Due to this, as shown in FIG. 1, low-pass filters 130 and 131 areprovided between the attenuators and the mixers 101 and 102,respectively. In FIG. 1, the outputs of the AD converters 402 and 403,and the noises included in signals reaching the IQ input terminals 108and 109 are the same as the example shown in FIG. 4. That is, the outputof the AD converter 402 contains a signal body 411, a turn noise 412,and a thermal noise 413 that the circuit has, as shown in therectangular 410 of FIG. 4. As indicated in the rectangle 414 of FIG. 4,signals in the IQ input terminals 108 and 109 contain the signal body415, and the noise 416 equal to or less than the cutoff frequencyf_(CUTOFF) of the external low-pass filters 404 and 405.

In the attenuators 103 and 104, similarly to the example shown in FIG.4, a wide band noise is generated, and the noise having a frequencyequal to or more than a cutoff frequency f_(CUTOFF2) is damped by alow-pass filter having the cutoff frequency f_(CUTOFF2) connectedimmediately before the mixers 101 and 102, as indicated in a rectangle118 of FIG. 1, and thereby the wide band noise caused by the attenuatorbecomes a noise 121 equal to or less than the cutoff frequencyf_(CUTOFF2). Thus, an output of the modulator 106 composed of the mixers101 and 102, as indicated in a rectangle 123, becomes a signal modulatedin both sides by regarding the carrier frequency “fc” as a center, butthe noise level becomes small which is a frequency equal to or more thanf_(CUTOUT2) distant from the center carrier frequency “fc” in themodulated RF signal. Therefore, it is possible to reduce the noise levelin receiving band without adding, to the RF output 132, an RF filter 430shown in FIG. 4.

FIG. 8 shows an example of a level diagram of a transmitter thatsatisfies the specification of a GSM1800. The transmitter is composed ofthe modulator 106, the driver amplifier 107, a power amplifier 801, anda front end circuit (FEM) 802 constituted by switches or the like. InFIG. 8, reference number 804 denotes an input end of the transmitter,and reference number 803 denotes an output end of the transmitter. Therespective gains of the driver amplifier 107, power amplifier 801, andfront end circuit 802, and performance of a noise index NF are estimatedfrom a circuit existing currently so as to have characteristics capableof being realized. In the case where an output signal level of themodulator 106 is −8 dBm, it is understood that the noise level inreceiving band at the input end 804 of the transmitter satisfies thespecification of the GSM1800 when being −160 dBm/Hz (−152 dBc) or less.

Specification of the case of the GSM is shown in FIG. 19, similarly toFIG. 8. As the frequency is lower, it is expected that each noise indexNF of the driver circuit 107 and the power amplifier 801 is reduced.However, since the specification is strict, it is necessary that thenoise level in receiving band is made −171 dBm/Hz (−166 dBc) or lesswhen the modulator 106 has an output of −5 dBm.

In order to make performance checks on a direct-up-conversiontransmitting circuit adopted in the present embodiment, the circuitshown in FIG. 5 has been manufactured for trial by a Bi-CMOS process inwhich a transistor cutoff frequency “f_(T)” is 20 GHz at a rule of 0.35μm, and evaluation thereof has been executed. The circuit manufacturedfor trial is a portion of a modulator 510 composed of: a group of IQsignal mixer circuits having the mixers 101 and 102; and bufferamplifiers 501 and 502 that amplify a mixer local signal being inputfrom an input terminal 505 of a 90° phase shifter 100. The 90° phaseshifter 100, the attenuators 508 and 509, and the low-pass filters 506and 507 are composed of individual parts. Each of the filters 506 and507 is a first order low-pass filter, and is composed of a resistor 511and a capacitor 512 as shown in the circuit diagram in a dotted linedenoted by reference number 500. A matching circuit 503 composed ofindividual parts is provided at an output end of the modulator 510, andis constituted to fetch a transmission signal from an output terminal504 after a 50Ω alignment and a differential-single conversion arecarried out.

In order to investigate effects of a low-pass filter in adirect-up-conversion transmitting circuit proposed in the presentembodiment, evaluation oriented to a GSM1800 has been carried out aboutthe case of no use of a filter and the case where the filter has cutofffrequencies of 4.9 MHz and 440 kHz. FIG. 6 shows evaluation resultsobtained when a local signal (LO) has an input level of 0 dBm, each IQinput signal has a DC level of 1.2 V, and an output has a frequency of1.75 GHz. An abscissa denotes a voltage level (dBV) of each IQ inputsignal, and a vertical axis denotes a ratio between a transmissionsignal and a noise in receiving band. In this figure, a characteristicline indicated by a black filled circle denotes noise characteristics inreceiving band, in the case of no use of filter. A characteristic lineindicated by a square denotes the case where a filter with a cutofffrequency of 440 kHz is used. A characteristic line indicated byasterisk “*” denotes noise characteristics in receiving band, in thecase where a filter with a cutoff frequency of 4.9 MHz is used. Thelevel of −17 dBV is an-allowable maximum input level to satisfy strainspecification included in a GSM, a GSM1800 and the like.

When the IQ input level is −17 dBV, an output thereof is −7 dBm. Incomparison with respective characteristics under this case, the case ofno use of filter has a level of −142 dBc/Hz (−149 dBm/Hz) while the caseof use of a filter having a cutoff frequency of 440 kHz has a level of−156 dBc/Hz (−163 dBm/Hz). Therefore, it is understood that the case ofuse of a filter having a cutoff frequency of 440 kHz is improved by 14dB. The similar results have been obtained even in the case of GSMtransmission frequency band. In this trial testing, althoughperformances relative to the GSM specification are insufficient, it isconsidered that improvement of device characteristics by use of a SiGe(silicon/germanium) bipolar transistor, and the like, can be achieved,and thereby effects of the present invention are expected.

Although it is possible to reduce the noise level in receiving band bylowering the cutoff frequency of the filter, degradation of phaseprecision of a modulation signal is considered due to an affect offrequency characteristics of a group delay. FIG. 7 shows a relationshipbetween the cutoff frequency of the filter and modulation signal ofphase precision. The GSM specification has a phase precision of 5° orless, but if a target value of the phase precision is set at 3° on thebasis of a margin thereof, sudden degradation is observed at the cutofffrequency of about 300 kHz or less. Therefore, a phase precision of 3°regarded as the target value cannot be satisfied. Assuming thatdispersion of resistance values of the integrated circuit is ±20% anddispersion of capacitance values thereof is ±30%, it is desired that adesign value of the cutoff frequency has lower limit of about 500 kHz.

As described above, according to the constitution of the firstembodiment that is the present invention in which a low-pass filter isconnected immediately before the IQ inputs of each mixer circuitconstituting a direct-up-conversion transmitting circuit, it isunderstood that the spur level in receiving band for an output can beeminently improved.

Second Embodiment

A second embodiment of the present invention will be described withreference to FIG. 9 and FIG. 10. In an example of the first embodimentdescribed above, there is shown characteristics of the case where thefirst order filter is connected immediately before the mixer. Althoughit is desirable that the cutoff frequency is lowered to reduce noise inreceiving band, such noise reduction is limited because the phaseprecision is affected by a group delay deviation which the filter has.Thus, in the present embodiment, an attempt is made to ensure dampingquantity in receiving band and suppression of the group delay deviationin signal band by using a high order filter.

FIG. 9A shows respective amplitude characteristics of a first orderlow-pass filter 90 a, a second order Butterworth filter 90 b, and athird order Butterworth filter 90 c. The characteristics of the firstorder low-pass filter in the cutoff frequency is 440 kHz, wherein thefirst order low-pass filter is set under the same conditions as thetesting results which is indicated by asterisks “*” shown in FIG. 6 andwhich is carried out for checks on the first embodiment. The dampingquantity at 20 MHz of the first order low-pass filter is about 33 dB.Each damping quantity of the second order and third order Butterworthfilters 90 b and 90 c is also set to have a value of 33 dB at 20 MHz.

FIG. 9B shows group delay characteristics of respective filters. The GSMsignal band is about 100 kHz. In the characteristics of the first orderfilter 90 a, a group delay deviation is 20 nsec within a band from 0 KHzto 100 KHz. An error of 1° in a signal of 100 kHz corresponds to 28nsec, and so, in the case of use of the first order filter 90 a, adeviation of about 0.7° occurs in a band of 100 kHz. Since spectrums ofthe GMSK modulation signal used in the GSM system are not uniform in aband of 100 kHz, this deviation is not equal to an absolute value ofphase precision of the signal. However, the case of no filter asindicated by the testing results shown in FIG. 7 is well coincident withdegenerative amount of phase errors of the case of using the first orderfilter having a cutoff frequency of 440 kHz. In the characteristics ofthe second order and third order Butterworth filters 90 b and 90 c, bothgroup delay deviations thereof are below 0.1 nsec, and so this does notaffect the phase deviation thereof. Therefore, in the case of changingfilter order from a first order to a second order or more, it is foundthat eminently improved effects are attained.

FIG. 10 shows a concrete example of a circuit constitution. The mixers101 and 102 are Gilbert type mixers widely known. Here, an I systemcircuit will be described in detail, and a description of a Q systemcircuit will be omitted because the Q system circuit is the same as theI system circuit. A local signal having a phase difference of 900 isgenerated at a phase shifter 1001 composed of a frequency divider. Afterthe local signal has been amplified at a buffer amplifier 501, twogroups of differential pairs 1006 which do switching operation in theGilbert mixer circuit 101 are driven. An I signal input 108 is damped atthe attenuator 103, and an input noise of 20 MHz or more is suppressedat a second order Sallen-Key type active low-pass filter 130 composed ofa feedback transistor 1003, a resistor 1004, and a capacitor 1005.

The Sallen-Key filter can be composed of a Butterworth type filter or aChabyshev type filter by selecting an element value. The filter outputdrives a mixer input stage transistor 1002, and is converted into a highfrequency signal by means of two groups of differential pairs 1006, andis fetched from respective connection ends between load resistances 1007and the differential pairs 1006 of a mixer. Here, although the secondorder filter is shown, the third order filter can easily be used insteadof the second order filter. However, as far as the GSM system isconcerned, as is evident from FIG. 9, necessary and sufficientcharacteristics can be obtained by the second order Butterworth typefilter. By the present embodiment, the direct-up-conversion transmittercapable of suppressing band free noises and reducing phase errors can beachieved.

Third Embodiment

A third embodiment of the present invention will be described withreference to FIG. 11. The same components as FIG. 1 are denoted by thesame reference number as FIG. 1. The present embodiment relates toreduction of a carrier leak generated by an effect of a DC offset of amixer circuit input in respective circuit constitutions of the first andsecond embodiments described above. In a direct-conversion transmittingcircuit that is the present invention, since many circuits such as afilter, a attenuator and the like are connected to a mixer input,increase in a DC offset generated at a mixer input terminal isconsidered. As a countermeasure thereof, a circuit constitution of thepresent embodiment is proposed.

First, a mixer carrier leak will be described here. This mixer functionsas a multiplier. As shown in formula (1), a modulation wave fc(t) isgenerated by multiplying a baseband input signal f(t) and a local signalcos(2πfc).fc(t)=f(t)×cos(2πfc)   (1)

When a DC offset a is added to the mixer input, as shown in formula (2),a single term of a carrier signal is generated and this causesdegradation of modulation precision.fc(t)=f(t)×cos(2πfc)+αcos(2πfc)   (2)

In order to correct the DC offset, in the present embodiment respectivechannels of the I and Q are provided with a bias correction circuit 1103consisting of an AD converter ADC for detecting an offset, a DAconverter DAC for generating a correction bias, and a control sectionCNT which carries out control for minimizing an offset andsimultaneously stores correction conditions. Correction is carried outwithin time from supply of power to beginning of transmission. Thecontrol section CNT is composed of a control register and a logiccircuit and the like. By the present embodiment, a direct-up-conversiontransmitter reducing an effect of a DC offset can be achieved.

Fourth Embodiment

A fourth embodiment according to the present invention will be describedwith reference to FIG. 12. The same components as FIG. 11 are denoted bythe same reference number as FIG. 1. In the third embodiment describedabove the DC offset correction circuit 1103 has been provided separatelyfrom the I and Q while in the present invention AD converters ADC whoseeach circuit scale become large are shared with the I and Q and therebythe entire circuit scales are reduced.

The AD converter ADC is selectively connected to the I and Q signallines by means of a switch SW. The DA converter DAC is providedexclusively for each of the I and Q signal lines. The control sectionCNT is also provided exclusively for each of the I and Q signal lines,and thereby each of the DC offsets is independently controlled. Sincecorrection cannot be made for the I and Q simultaneously, correctiontime of the present embodiment is required about twice further than thatof the third embodiment. By the present embodiment, adirect-up-conversion transmitter reducing an effect of the DC offset canbe achieved with a small circuit scale.

Fifth Embodiment

A fifth embodiment according to the present invention will be describedwith reference to FIG. 13. The same components as FIG. 10 are denoted bythe same reference number as FIG. 10. In the second embodiment shown inFIG. 10, the active low-pass filter 130 and the mixer 101 have beenconstituted as independent circuits, respectively. In contrast, in thepresent embodiment, respective functions of both the input transistor1002 of the mixer and the emitter follower circuit transistor 1003 of anactive low-pass filter 130, as shown in FIG. 10, are integrated with atransistor 1302 shown in FIG. 13. In this manner, it is possible toachieve a circuit which is not saturated even if a large baseband signalis applied thereto.

The I input signal 108 is converted to current from voltage at adifferential input circuit composed of a PNP type transistor 1303. Thetransistors 1301 and 1302 each have a current mirror structure, andcurrent is returned to two groups of differential pairs 1006 for mixer,and a mixer output is supplied from a connection end connected to eachload resistor 1007. A low-pass filter 1300 is composed of resistors R1and R2 connected in series to bases of the transistors 1301 and 1302,and capacitors C1 and C2 connected to a grounding terminal, and anemitter of the transistor 1302, respectively. In addition, emitters ofthe transistors 1301 and 1302 are grounded via resistors R3 and R4,respectively. By the present embodiment, it is possible to achieve adirect-conversion transmitter capable of corresponding to a largebaseband input signal.

Sixth Embodiment

A sixth embodiment according to the present invention will be describedwith reference to FIG. 14. The same components as FIG. 10 are denoted bythe same reference number as FIG. 10. In the fifth embodiment shown inFIG. 13, since the resistors R1 and R2 each constituting the filter areconnected in series to bases of the transistors 1301 and 1302, increasein DC offset of a mixer input voltage is considered due to deviation ofthe base current caused by a base/collector current amplification rate“hFE” of each transistor.

In contrast, in the present embodiment, MOSFETs 1400 and 1401 eachhaving a gate in which no DC current flows are applied to a currentmirror section, and thereby generation of a DC offset caused bydeviation of potential fall at the resistors R1 and R2 is suppressed. Inaddition, a transistor constituting the attenuator 103 is changed fromthe PNP transistor 1303 to a P type MOSFET 1402. This is because inputimpedances thereof are increased and driving thereof can be achieved byusing small amount of power.

By the present embodiment, since a large resistor can be applied to afilter, capacitive value thereof can be reduced. As a result, a lownoise direct-conversion transmitter having small element area can beachieved.

Seventh Embodiment

A seventh embodiment according to the present invention will bedescribed with reference to FIG. 15. The present embodiment is atransceiver IC adopting direct-conversion and usingtransmission/reception that is applied to a triple band ofGSM/GSM1800/GSM1900. A receiving circuit of this transceiver IC 150receives frequency bands from 925 to 960 MHz, 1805 to 1880 MHz, and 1930to 1990 MHz of the respective GSM/GSM 1800/GSM1900. A large interferencewave other than each frequency band is erased by means of an external RFfilter 1506. Therefore, the signal is amplified by low noise amplifiers1507, 1508 and 1509 extensively provided relative to the respectivefrequency bands. Respective outputs of the low noise amplifiers areconnected to an input of common direct-conversion mixers 1510 and 1511operating in all the frequency bands, and the signals are directlyconverted into I and Q components of a baseband frequency. The mixers1510 and 1511 are driven by means of local signals each having a 90°phase difference generated by a ½ frequency divider 1512. The signalhaving I and Q components are subjected to processing for eliminatinginterference waves and for gain regulation in baseband programmable gainvariable amplifiers/channel low-pass filter rows (PGA & LPS) 1513 and1514, and thereafter is output as I and Q signals in a downstream sideof the circuit.

A transmitting circuit applies any of the embodiments introducedpreviously. The IQ transmission signals are adjusted to a desired signallevel at the attenuator 103, and a wide band noise generated by theattenuator 103 is suppressed at the low-pass filter 130. The signalswhose noises are suppressed at the filters 130 are converted intomodulation signals having RF frequency, by the modulator 106 composed ofa group of mixers 101 and 102. These mixers operate within respectiveranges from 880 to 915 MHz in the case of GSM, from 1710 to 1785 MHz inthe case of the GSM1800, and from 1850 to 1910 MHz in the case of theGSM1900. The mixers 101 and 102 each are driven by means of a localsignal having a 90° phase difference generated by the frequency divider100.

The output signal of the modulator 106 is amplified by a GSM driveramplification circuit 1500 or a driver circuit 1501 compatible with theGSM1800 and GSM1900. A band pass filter 1502 such as a SAW filter havingrapid waveform characteristics, or the like is connected to an output ofthe GSM driver circuit, and thereby residual noise in receiving bandwhich is 20 MHz distant is eliminated. Here, although the SAW filterhaving rapid waveform characteristics is connected to the GSM output inaccordance with the testing results shown in the first embodiment, thisfilter can replace an inexpensive LC filter according to higherperformance of the circuit. The output signal of the filter is amplifiedby a power amplifier module (PA module) 801.

A simple LC filter 1503 is connected to an output of the driveramplification circuit 1501 compatible with the GSM1800 and GSM1900, andthe signal thereof is amplified by the power amplifier module 801 afterhigh harmonics are eliminated. Here, the power amplifier module 801packages the GSM modulator and the modulator compatible with the GSM1800and GSM1900. The amplified signal is transmitted from an antenna via alow-pass filter (LPF) 1504 that eliminates high harmonics generated byan output of the amplifier in the power amplifier module 801, and via atransmission/reception changeover switch (S/W) 1505.

A voltage control oscillator (RF VCO) 1515 receives and constantlyoscillates a feedback loop by means of a synthesizer (RF PLL Synth)1516, and generates a transmission/reception signal as follows.

GSM reception: An oscillator 1515 oscillates within a range from 3700 to3840 MHz. The output of this oscillator is frequency-divided into twosections by means of a frequency divider 1517, and further isfrequency-divided into two sections by the frequency divider 1512.Thereby, a local signal for a GSM reception, which drives the mixers1510 and 1511, is obtained.

GSM1800 reception: The oscillator 1515 oscillates within a range from3610 to 3760 MHz. The output of this oscillator is directly connected tothe frequency divider 1512 without passing through the frequency divider1517, and is frequency-divided into two sections by a switch 1518.Thereby, the local signal for the GSM1800 reception, which drives themixers 1510 and 1511, is obtained.

GSM1900 reception: The oscillator 1515 oscillates within a range from3860 to 3980 MHz. The output of this oscillator is directly connected tothe frequency divider 1512 without passing through the frequency divider1517, and is frequency-divided into two section by the switch 1518.Thereby, the local signal for the GSM1900 reception, which drives themixers 1510 and 1511, is obtained.

GSM transmission: The oscillator 1515 oscillates within a range from3520 to 3660 MHz. The output of this oscillator is frequency-dividedinto two sections by means of a frequency divider 1519, and further isfrequency-divided by means of the frequency divider 100. Thereby, alocal signal for the GSM transmission, which drives the modulator 106,is obtained.

GSM1800 transmission: The oscillator 1515 oscillates within a range from3420 to 3570 MHz. The output of this oscillator is directly connected tothe frequency divider 100 without passing through the frequency divider1519, and is frequency-divided into two sections by means of a switch1520 without passing through a frequency divider 1519. Thereby, a localsignal for the GSM1800 transmission, which drives the modulator 106, isobtained.

GSM1900 transmission: The oscillator 1515 oscillates within a range from3700 to 3820 MHz. The output of this oscillator is directly connected tothe frequency divider 100 without passing through the frequency divider1519, and is frequency-divided into two sections by the switch 1520.Thereby, a local signal for the GSM1900 transmission, which drives themodulator 106, is obtained.

In order to make such operations, the oscillator 1515 operates within arange from 3420 to 3980 MHz. By the present embodiment, adirect-conversion circuit can be achieved for both of transmission andreception by using one voltage control oscillator.

Eighth Embodiment

An eighth embodiment according to the present invention will bedescribed with reference to FIG. 16 and FIG. 220. In FIG. 16, the samecomponents as FIG. 15 are denoted by the same reference number as FIG.15. In the sixth embodiment described previously, the case of the GSMfrequency-divides the output of the oscillator 1515 into four sections,and the cases of the GSM1800 and the GSM1900 each are frequency-dividedinto four sections and generate the local signal. FIG. 20 shows an inputwaveform, a two-frequency division waveform, and a four-frequencywaveform of a two-frequency divider (i.e., an output waveform of theoscillator 1515).

When the output of the oscillator 1515 is inputted into thetwo-frequency divider, two waveforms of outputs 1 and 2 are generated.One of two rising edges 2007 and 2008 in the waveforms of the outputs 1and 2 is synchronized with a rising edges 2005 of the input of thefrequency divider (that is, the output of the oscillator), and the otheris synchronized with a falling edge 2006 of the input of the frequencydivider.

If the output of the oscillator 1515 has a duty ratio of 50%, a phasedifference between these two outputs is 90°. In the case where the dutyratio is shifted from 50%, an error occurs in the phase difference. Whenthe waveform of the output 1 is further frequency-divided into twosections, waveforms of outputs 3 and 4 are obtained. Rising edges 2009and 2010 of any of the waveforms are also synchronized with the risingedge 2005 of the oscillator output, and a signal precisely having aphase difference of 90° phase difference can be generated withoutdepending on the duty ratio of the oscillation waveforms.

Therefore, although a signal having a precise phase difference can begenerated relative to the GSM in the sixth embodiment described above,there occurs each error depending on the duty ratio of the oscillationwaveforms in the GSM1800 and the GSM1900. The transceiver IC 160employing a direct-conversion for both of transmission and reception,which is applied to the triple band of the GSM/GSM1800/GSM1900 in thepresent embodiment shown in FIG. 16, is constituted in which theoscillation frequency of the voltage control oscillator 1515 is setwithin such a range from 6840 to 7960 MHz as to double the sixthembodiment, and a frequency divider 1600 is newly connected to theoscillator output. In this manner, by using eight-frequency division forthe GSM and using four-frequency divisions for the GSM1800 and GSM1900,a local signal precisely having a phase difference of 90° can begenerated without depending on the duty ratio of the oscillatorwaveforms.

Ninth Embodiment

A ninth embodiment according to the present invention will be describedwith reference to FIG. 17 and FIG. 19. In FIG. 17, the same componentsas FIG. 15 are denoted by the same reference number as FIG. 15. Thesixth and seventh embodiments described above has been the case of thetriple band compatible ICs for the GSM, GSM1800 and GSM1900. Incontrast, the present embodiment is the case a 4-band compatibletransceiver IC to which a GMS850 is newly added. A low noise amplifier1700 for receiving a range from 869 to 894 MHz has been added to areceiving circuit of this transceiver IC 170. In the transmittingcircuit, a GSM exclusive driver circuit 1500 is actuated for signals ofboth of the GSM and GSM850. External RF filters for the GSM and GSM850each are composed of an LC filter, and eliminates high harmonics. Inorder to use the GSM and GSM850 circuits in combination, during the GSMand GSM850 transmissions, the noise level in receiving band for thedriver circuit output is required to satisfy a value of −160 dBm/Hz orless at an output of 3 dBm (see FIG. 19). In the case where the noiselevel does not meet this condition, there may be provided a constructionin which the driver circuits 1500 compatible currently therewith areprovided for each of the GSM and GSM850 to increases two systems, and anexclusive SAW filter is used for each of the systems.

By the present embodiment described above, a 4-band compatibletransceiver IC can be achieved by using the small number of externalelements.

As described above, several preferred embodiments of the presentinvention has been described. However, the present invention is notlimited to these embodiments, and of course various design modificationsthereof can be made without departing from the spirit of the presentinvention.

According to the present invention, in comparison with a conventionaltransmitter applying offset PLL, even if a required external VCO inaddition to an RF integrated circuit, a power amplifier and a front endcircuit is reduced and current transistor performance is maintained,then a GSM/GSM1800/GSM 1900 triple band transmitter/receiver can beachieved by using one filter having rapid waveform characteristics, suchas a SAW or the like more inexpensive than the VCO. Further, byimproving transistor characteristics, a triple band or quadrant bandtransmitter/receiver can be formed without using expensive externalparts.

1. A direct-conversion transmitting circuit comprising: a modulator tomodulate I and Q signals into a transmitting frequency signal, the I andQ signals being inputted from a base band circuit to said modulator; afirst driver amplification circuit coupled to an output node of saidmodulator to amplify a first transmitting frequency signal beingmodulated into a first frequency band through said modulator; a seconddriver amplification circuit coupled to an output node of said modulatorto amplify a second transmitting frequency signal being modulated into asecond frequency band through said modulator, the second frequency bandbeing higher than the first frequency band; first and second low-passfilters being coupled at output nodes thereof to input nodes of saidmodulator; first and second gain/bias adjusters being coupled at outputnodes thereof to input nodes of said first and second low-pass filters,respectively; and wherein said modulator comprises first and secondmixers, and a first phase shifter, wherein high frequency outputterminals of said first and second mixers are connected to each other,wherein an output terminal of said first low-pass filter is connected toan input terminal of said first mixer, and an input terminal of saidfirst low-pass filter is connected to an output terminal of said firstgain/bias adjuster to suppress a noise generated by said first gain/biasadjuster, wherein an output terminal of said second low-pass filter isconnected to an input terminal of said second mixer, and an inputterminal of said second low-pass filter is connected to an outputterminal of said second gain/bias adjuster to suppress a noise generatedby said second gain/bias adjuster, wherein a first output terminal ofsaid first phase shifter is connected to a local signal input terminalof said first mixer, and a second output terminal of said first phaseshifter is connected to a local signal input terminal of said secondmixer, wherein an input signal generated from an output signal of afirst AD converter is applied to an input terminal of said firstgain/bias adjuster to reduce difference in gain and bias levels betweenan input signal of said first mixer and an output signal of said firstAD converter, and wherein an input signal generated from an outputsignal of a second AD converter is applied to an input terminal of saidsecond gain/bias adjuster to reduce difference in gain and bias levelsbetween an input signal of said second mixer and an output signal ofsaid second AD converter.
 2. The direct-conversion transmitting circuitaccording to claim 1, wherein said first phase shifter is comprised of afrequency divider circuit.
 3. The direct-conversion transmitting circuitaccording to claim 1, wherein each circuit of said first and secondlow-pass filters is comprised of a filter whose order is at least asecond order.